This invention relates to telephony transmission systems and, more particularly, to a carrier supply for a frequency division multiplexed system.
In modern telephony frequency division multiplex systems, the final step of the modulation process occurs when the message channels are modulated at relatively high frequencies for transmission over the selected medium. A simplified block diagram embodiment of the prior art circuitry used for this final step of modulation in a frequency division multiplexed system is shown in FIG. 1. In the transmitting terminal of FIG. 1, the input signal to the terminal, S, would pass through the input low-pass filter to the first stage of modulation where the input signal is modulated at the carrier frequency f.sub.1. (For illustrative purposes, lower single sideband modulation is employed throughout FIG. 1 as can be seen from the designations thereon indicating the numerical value of the frequencies at various points.) The modulating frequency f.sub.1 is supplied by an oscillator which inherently has a relatively small output frequency variation designated .+-..DELTA.f.sub.1 in the drawing. The output of the first oscillator is then fed to an input of a summing network, the other input of which is connected to the output of the carrier oscillator connected to the first modulator. The combined output of the summing network is then fed to a second modulator which is connected to a second carrier oscillator having an output frequency f.sub.2 and an inherent output frequency variation designated as .+-..DELTA.f.sub.2. The output carrier pilot frequency of this second stage of modulation is nominally the numerical difference of the frequency of the second oscillator and the first oscillator as indicated on the drawing. For example, if f.sub.1 were chosen to nominally be 5.622 mHz and f.sub.2 as nominally 11.8 mHz then the carrier pilot frequency would nominally be 11.8-5.622=6.178 mHz. This difference frequency is then transmitted through a bandpass filter along with the modulated input signal, S, and combined with other channels for transmission over a medium, such as a coaxial cable, to a compatible demodulating terminal.
The typical prior art demodulator illustrated in FIG. 1 has an input bandpass filter which selects from a decombiner circuit the particular signal transmitted by the transmitter illustrated in FIG. 1. An oscillator connected to the first demodulator has an output frequency f.sub.2, which is normally equal to the frequency f.sub.2 in the transmitter, and a small output frequency variation designated as .+-..DELTA.f.sub.2. (Although for purposes of continuity in the discussion the designation .DELTA.f.sub.2 is used to represent the frequency variation of comparable oscillators in the transmitter and receiver, the output frequency variations of these oscillators would generally not be the same at any given instant of time.) The output of this first demodulator is then fed through a bandpass filter to a hybrid network and ultimately a second demodulator. A phase detector is normally connected to both the hybrid network to be responsive to the phase and frequency of the signal demodulated by the first demodulator and to a voltage controlled oscillator having an output frequency f.sub.1 with an output variation .+-..DELTA.f.sub.1. The phase detector and oscillator are interconnected in an automatic phase control (APC) loop to provide compensation for the phase and frequency variations in both transmitter oscillators and the oscillator connected to the first demodulator in the receiver, as well as the phase and frequency variations suffered during transmission over the transmission medium. The output of the second demodulator is fed through the low-pass filter to the following circuitry.
The frequency variation error or offset in a system such as that illustrated in FIG. 1 is thus the combination of the variations in both oscillators in the transmitter and the oscillator connected to the first demodulator in the receiver. The "worst case" frequency offset condition as indicated by the notations of FIG. 1 would then be the sum (.DELTA.f.sub.1 +.DELTA.f.sub.2 +.DELTA.f.sub.2) of the maximum variations of each of the noted oscillators. This relatively large frequency offset implies increased quadrature distortion and phase error, both of which result in signal degradation. As the frequency offset requirements of modern transmission systems with wider bandwidths become more and more stringent, it becomes increasingly difficult from a cost and reliability standpoint to design carrier supplies with the required frequency stability. For example, a typical frequency offset requirement for a modern analog transmission system having carrier frequencies of up to 70 mHz might be 2 Hz or less. For digital transmission in such a system, the frequency offset requirement is as near zero as practical.
The need for stable phase and frequency synchronized carrier supplies has thus grown considerably in recent years. In these supplies, carrier synchronization is often accomplished by locking the phase of the carrier frequency in the receiver to a synchronizing pilot that is transmitted along with the broadband signal. As illustrated in the prior art system of FIG. 1, an automatic phase control circuit using a voltage controlled oscillator is commonly employed for this purpose. The steady state phase error of the automatic phase control circuit, which as noted heretofore introduces signal degradation, is directly related to the magnitude of the frequency offset. As can be seen from the frequency notations of FIG. 1, the frequency offset of such prior art systems is relatively high. As also noted heretofore, the frequency offset is mainly due to the output frequency variations of the carrier oscillators.
In a system such as that of FIG. 1 several straightforward approaches may be taken to reduce the frequency offset and thereby minimize the signal degradation. For example, the oscillators may be crystal oscillators with only a minimal frequency variation; the gain of the automatic phase control loop could be increased to reduce the phase error; and two automatic phase control loops could be employed in the receiver, one for each demodulator. These procedures, however, have generally not reduced the frequency offset to levels sufficient for modern high frequency wideband systems. Expensive and complex crystal oscillators have been designed to reduce the frequency offset to less than 10 cycles, which is sufficient for the narrower band systems of the past but is unacceptable from a cost and design standpoint for modern systems. Use of two automatic phase control loops introduces difficult circuit design problems and, in all probability, resultant circuitry which is also complex and expensive, while increased loop gain to the necessary levels in a single automatic phase control loop introduces all the problems and errors incurred with high gain loops without obtaining the desired results for modern high frequency systems. Such automatic phase control loops are additionally difficult to design because of the wider frequency capture range these circuits must have for higher frequency offsets.
It is, therefore, an object of this invention to provide a wideband carrier system having a minimal transmitted frequency offset equivalent to that of only a single carrier oscillator.
It is another object of this invention to provide a modern wideband carrier system which may employ relatively simple oscillators and only a single, relatively low gain, automatic phase control loop.